1. Field of the Invention
The present invention relates to output stages of amplifiers. More particularly, the present invention relates to methods and apparatus for setting and controlling quiescent (bias) current and cross over distortion independently in amplifier output stages. In addition, the present invention relates to amplifier output stages with inherent short circuit protection.
2. Description of the Related Art
An output stage, as the final stage of amplifiers, generally deals with relatively large signals. The most challenging requirement in the design of the output stage is that it deliver the required amount of power to the load in an efficient manner. This implies that the power dissipated in the output stage transistors must be as low as possible. This is primarily due to the fact that the power dissipated in a transistor raises its internal junction temperature, and there is a maximum temperature (in the range of 150.degree. C. to 200.degree. C. for silicon devices) above which the transistor is destroyed. Other reasons for requiring high-power conversion efficiency are to prolong the life of the batteries used in battery-powered circuits, to permit a smaller, lower-cost power supply, or to obviate the need for cooling fans.
The output stage of an amplifier is generally designed to deliver a substantial amount of power into a low-impedance load with acceptably low levels of signal distortion. Therefore, the amplifier output stages are generally designed to have one or more of the following properties: large output current swing, large output voltage swing, low output impedance, and low standby power. Additionally, the output stage is further designed to have sufficiently good frequency response such that it will not present a limitation to the rest of the amplifier circuit.
The output stage power transistors are generally several hundred to thousand times bigger than the diode connected transistors that is used for their quiescent current setting, and thus, their base-emitter voltage V.sub.BE is different. Therefore, emitter degeneration resistors are used in both output stage power transistors and diode connected transistors to help in setting the quiescent current. However, as the temperature of output stage power transistors rises, the quiescent current changes as a result since the diode connected transistors are not at the same temperature as the output stage power transistors. Therefore, setting and controlling a temperature stable quiescent current has been a major challenge.
One conventional approach to this problem is through several mask changes to properly adjust the quiescent current. However, the quiescent current adjusted through several mask changes still exhibit variations among the different wafers due to process tolerances. Moreover, several mask changes to properly adjust the quiescent current takes significant amount of time, substantially prolonging the time to market period.
Furthermore, one of the major failures of these output stages is the possible burn-out of the output devices or interconnections due to accidental or transient overload conditions given that the output stages are designed to handle large signal swings at relatively high current levels. A typical example of such an overload condition would be the accidental short circuit of the output to the power supply or the ground terminals. In such cases, the burn-out of the output devices or other permanent damage to the monolithic circuit can be avoided by limiting the maximum available output current to a safe value. This safe value is determined by the size and layout of the output devices and by the maximum allowable power dissipation considerations.
FIG. 1 shows a conventional class AB output stage with short circuit protection which operates in the event of an output short circuit while the output voltage V.sub.OUT is positive. As shown, two resistors R.sub.E1 and R.sub.E2 in series are coupled between the emitter terminal of transistor T.sub.1 and the emitter terminal of transistor T.sub.2. Furthermore, the base terminal of transistor T.sub.3 is coupled to the emitter terminal of transistor T.sub.1, while the base terminal of transistor T.sub.1 is coupled to the collector terminal of transistor T.sub.3. As further shown in FIG. 1, the emitter terminal of transistor T.sub.3 is coupled to the output node of the class AB output stage. Also shown in FIG. 1 are two diodes D.sub.1 and D.sub.2 coupled in series between the collector terminal of transistor T.sub.3 and the base terminal of transistor T.sub.2.
Accordingly, in the manner described above, a large current which flows through transistor T.sub.1 in the event of a short circuit will develop a voltage across resistor R.sub.E1 having a sufficient value to turn transistor T.sub.3 on. The collector of transistor T.sub.3 will then conduct most of the bias current I.sub.BIAS taking transistor T.sub.1 out of its base drive. The current through transistor T.sub.1 will therefore be reduced to a safe operating level. Thus, this short circuit protection approach ensures device safety.
However, the above short circuit protection approach has the disadvantage that under normal operating conditions, about 0.5 volt drop may appear across each resistors R.sub.E1 and R.sub.E2. This translates into an equivalent reduction in voltage swing at the output in each direction, thus resulting in reduction in efficiency.
FIG. 2 shows a conventional thermal shutdown circuit. As shown, a zener diode Z.sub.1 is coupled between the base terminal of transistor T.sub.1 and the emitter terminal of a turn-on transistor T.sub.2. Transistor T.sub.2 is normally configured to be off. As the chip temperature rises, the combination of the positive temperature coefficient of zener diode Z.sub.1 and the negative temperature coefficient of transistor T.sub.1 base-emitter voltage V.sub.BE1 cancels each other such that the emitter voltage V.sub.E1 of transistor T.sub.1 remains unchanged.
Since the base voltage V.sub.B of transistor T.sub.2 is obtained from the emitter voltage V.sub.E1 of transistor T.sub.1 with a resistor divider (comprising resistors R.sub.1 and R.sub.2), the base voltage V.sub.B of the of transistor T.sub.2 likewise remains unchanged. On the other hand, the base-emitter voltage V.sub.BE2 of transistor T.sub.2 decreases with the increase in temperature at a rate of approximately 2 mV/.degree. C. and transistor T.sub.2 begins conducting. Naturally, the precise point at which transistor T.sub.2 begins conducting would largely depend upon the parameters of the transistor T.sub.2 itself. Thus, by setting the temperature of the base voltage V.sub.B of transistor T.sub.2 at 150.degree. C., thermal shutdown can be achieved.
In this manner, the turn-on transistor T.sub.2 is configured to absorb the bias current of the amplifier, therefore virtually shutting down its operation in the event that the temperature exceeds a safe preset value.
The conventional output stages are generally voltage drivers and therefore lack inherent current limitation. To cure this, the output stages require some type of feedback for short circuit protection. The traditional method uses a resistor across the emitter-base junction of the short circuit protection transistor to turn it on once a threshold load current is passed. This, in turn, consumes the drive current of the output stage and therefore shuts it down. This approach, however, is not very precise since the exact value of the short circuit protection transistor base-emitter voltage V.sub.BE is not known at the elevated temperature beyond the safe preset value, and where the value of the resistor is too small to be accurately manufactured. For example, a resistor having a value of 0.25 Ohms is necessary for a threshold current of 2 amperes and a base-emitter voltage V.sub.BE of the short circuit protection transistor at 0.5 Volts.
In accordance with another approach, the output stage power transistor base current I.sub.B is used along with a complicated circuitry to perform the short circuit protection. However, since this latter approach involves the .beta. of the power transistor, the total accuracy (for example, a desired accuracy of .+-.20% for a short circuit current of 2 A) protection in this manner significantly decreases. More particularly, since the base current I.sub.B of a bipolar transistor is approximately equal to the emitter terminal current I.sub.E divided by the .beta., the variation in the bipolar transistor emitter current I.sub.E is correspondingly reflected in the base terminal current I.sub.B. Therefore, the .beta. of the output stage power transistor significantly decreases the total accuracy since the .beta. for a bipolar transistor varies greatly (for example, from 30 to 200) depending upon the specific wafer.